Recently, increases in the speed and complexity of integrated circuits have worsened the problems of operation failure, a decrease in operating band, and the like due to noise in chips, variations in chip noise, temporal fluctuations in clock phase (jitter), and the like. Conventionally, signals are extracted outside an integrated circuit, and the behaviors of the signals are observed. Although the operating frequencies of integrated circuits have increased year by year, the operating frequencies of boards and packages outside the integrated circuits have not followed the increasing speeds. It is therefore difficult to observe high-speed operation in integrated circuits on the outside of the integrated circuits. Therefore, greater importance has been attached to a technique of observing internal operation by providing a measuring circuit in a chip.
FIG. 17 shows a conventional measuring circuit provided in a chip. A measuring circuit 1801 is driven by a sampling clock 1804 to operate at every period T. More specifically, this circuit compares the value of a to-be-measured signal 1802 with the value (m×Td) of a reference signal 1803 to obtain m by which the magnitude relation between the to-be-measured signal 1802 and the reference signal 1803 is reversed. The circuit then sets m as an output 1805 to estimate that the value of the to-be-measured signal 1802 is (m×Td). In this case, m represents an integer, and Td represents reference value intervals, i.e., a resolution.
This method uses analog/digital conversion to convert the to-be-measured signal 1802 which is a continuous value into the digital value m, and hence need not output any analog signal which is susceptible to noise and difficult to measure. A problem in this method is that errors (quantization errors) corresponding to Td at maximum occur between values 1904, 1905, and 1906 (m×Td+Δa, (m+1)×Td+Δb, and m×Td+Δc) of to-be-measured signals and measurement results (m, m+1, and m), as shown in FIG. 18.
To-be-measured signals include two types of signals, i.e., a continuous signal and a discrete signal. There are different noise countermeasures for the respective measurements. A continuous signal is a continuous value like noise on a power supply line. A discrete signal is a value generated only once at the period T like that of the timing fluctuation (jitter) of a clock.
For the measurement of continuous signals, there is available an oversampling measurement technique of using an array of N measuring circuits 2002 and setting the operation timings of them to N-phase clocks CK1 to CKN with the period T. In this case, since the N measuring circuits 2002 can detect the value of to-be-measured signals 2001 at every time T/N, the sampling frequency becomes N times higher than that in the case in which one measuring circuit is used.
FIG. 20 shows sampling timings when oversampling is performed. FIG. 21 shows sampling timings when no oversampling is performed. Reference numeral 2101 denotes a to-be-measured signal; and 2102 to 2104, sampling timings. Reference symbols D1 to D4 denote sampling timings.
FIG. 22 shows the frequency spectrum converted from the measurement result in FIG. 20. FIG. 23 shows the frequency spectrum converted from the measurement result in FIG. 21. Reference numerals 301 and 401 denote the spectra of signals; 302 and 402, noise components; and 304, the cutoff characteristic of a low-pass filter. Reference symbol Fck denotes the operating frequency (1/T) of a signal.
As is obvious from the sampling theorem, using the above oversampling processing makes it possible to measure up to a high-frequency component N times higher in frequency than that in the case in which one measuring circuit is used. With oversampling processing, although the band of a noise component, in particular, extends to a high frequency, the intensity of the component per unit frequency decreases. As shown in FIG. 22, therefore, the signal-to-noise ratio can be improved by cutting off the high-frequency component of the noise component 302 using a low-pass filter.
According to another measuring method, as shown in FIG. 24, using a measuring circuit 2301 capable of operating at a period T/N and a sampling clock 2303 with the period T/N allows one measuring circuit 2301 to implement the same measurement as that described above. Reference numeral 2302 denotes a to-be-measured signal; and 2304, an output.
An example of a discrete signal is clock jitter J like that shown in FIG. 25, which indicates how much the phase of a to-be-measured signal 2402 shifts from that of a reference signal 2401 at maximum.
A method of measuring this clock jitter is disclosed in, for example, Japanese Patent Laid-Open No. 2000-111587 (reference 1). As shown in FIG. 26, a to-be-measured signal SIG is made to pass through a plurality of delay elements 2501 each having a delay value Ts to shift the phase of the to-be-measured signal SIG by Ts at a time. Flip-flops 2502, 2503, 2504, . . . then compare these phases of the to-be-measured signal SIG with a reference signal REF. If the to-be-measured signal SIG is shifted from the reference signal REF by a time Tjit, the flip-flops 2502 and 2503 respectively receive phase differences Tjit and Tjit−Ts. Each of the flip-flops 2502, 2503, 2504, . . . is a circuit which outputs “1” when the phase difference is 0 or more, and outputs “0” otherwise. If a phase difference Tjit−N×Ts is 0 or less, the (N+1)th flip-flop outputs “0” for the first time. Observing this result will determine the value of N. Outputting the value N makes it possible to measure that Tjit is about N×Ts.
In this method, as shown in FIG. 27, errors (quantization errors) corresponding to Td at maximum occur between phase differences 804, 805, and 806 (m×Td+Δa, (m+1)×Td+Δb, and m×Td+Δc) between the reference signal REF and the to-be-measured signal SIG and measurement results (m, m+1, and m). Such an error component becomes quantization noise, which is a cause of deterioration in signal-to-noise ratio.
In addition, as shown in FIG. 28, this method performs one measurement at every period T of the reference signal REF and to-be-measured signal SIG to obtain phase differences Δ1, Δ2, Δ3, . . . FIG. 29 shows this jitter value information in a spectral form, in which a signal component 1001 and a quantization noise component 1002 occur in a frequency band equal to or less than Fck/2 which is half a sampling frequency Fck (equal to a clock frequency 1/T of the to-be-measured signal) (the quantization noise component 1002 uniformly exists in the band of a frequency of 0 to a frequency of Fck/2). If a low-pass filter having the cutoff characteristic indicated by a dotted line 304 is used to remove the quantization noise component 1002, the filter degrades the signal component 1001 as well. This makes it impossible to use the filter, and hence quantization noise cannot be reduced.
Unlike in a continuous signal, in a discrete signal, the interval of a to-be-measured signal does not exist in the period T or less. The oversampling technique, which measures a to-be-measured signal at a period shorter than the period T, cannot be directly applied to a discrete signal. In order to reduce noise due to an error, it is, therefore, necessary to design a measuring instrument which can reduce a quantization error Ts.
In general, however, as the measurement resolution of a measuring instrument increases (Ts decreases), the operation of the instrument becomes complicated, and more outputs are required. In addition, the operation speed of the measuring instrument has its upper limit, and the frequency band which allows output signals to be extracted to the outside of the integrated circuit also has its upper limit. It is therefore very difficult to achieve a high resolution by using the conventional methods.